Array Antenna System and Spread Spectrum Beamformer Method

ABSTRACT

A method for transmitting digital beamformed signals in a transmit array antenna apparatus utilizing a single transceiver with one power amplifier, one up-frequency converter and one digital-to-analog converter for said array transmit antenna apparatus comprising the steps of: generating a first set of direct-sequence spread spectrum codes; generating a plurality of weights, each weight being a beamforming amplitude and phase or delay for each element; generating a direct-sequence spread spectrum multiplexed signal containing such weights while using one of such first-set codes per element; converting such an multiplexed signal to a convenient radio frequency; amplifying and transmitting such a multiplexed radio frequency signal to the elements; generating a second set of direct-sequence spread spectrum codes; extracting a radio frequency signal with direction-bearing weight information at each element while using a subset of codes from the second set; generating a third set of direct-sequence spread spectrum codes at each element; transmitting a signal with array gain beamformed towards a specific direction while using a transmit array apparatus composed of spaced elements, such a transmit beamformed signal being a radio frequency signal, a direct-sequence spread spectrum radio frequency signal containing a subset of codes from the third set, or a sequence of radio frequency pulses that have short duration and high power.

FIELD OF THE INVENTION

The present invention relates generally to array antennas with digital beamformers and their use in radar, sensor and communication applications, and particularly to radar, sensor and communication systems using direct sequence spread spectrum signaling and array antennas with digital beamformers using spread spectrum multiplexing within the array system. The apparatus and methods of this invention include methods for transmitting and receiving signals with antenna array gain and methods, devices and techniques for multiplexing the signals within the array apparatus.

BACKGROUND OF THE INVENTION Prior Art Antenna Array with Digitally Controlled Analog Beamformer

FIG. 1 illustrates a prior-art exemplary receive array 100 with digitally controlled analog beamformer 190. Typically, the phase shifters at each antenna have high insertion loss and must be preceded by a low-noise amplifier (LNA). The system includes an antenna array 110 with an arbitrary number M of elements 111-119, each element with a dedicated LNA 121-129 and phase-shifter 131-139, and each phase shifter with electronics and interfaces for digital control and calibration. The amplitude and/or phase weights are generated at a control processor 150 and transmitted digitally to the control interface of the phase shifter at each elemental antenna. Actual beamforming occurs at RF—prior to demodulation—as the signals received by each element are added together in the Analog Summation Network 140. The resulting RF signal is then converted to baseband using a down-converter, and then to a discrete-time signal using an analog-to-digital converter and digital receiver 170.

FIG. 11 illustrates a prior-art exemplary transmit array 1100 with digitally controlled analog beamformer 1190. The system includes an antenna array 1110 with an arbitrary number M of elements 1111-1119, each element with a dedicated RF Power Amplifier (PA) 1121-1129 and phase-shifter 1131-1139, and each phase shifter with electronics and interfaces for digital control and calibration. The amplitude and/or phase weights are generated at a control processor 1150 and transmitted digitally to the control interface of the phase shifter at each elemental antenna. The baseband signal 1191 is converted to analog using a digital transmitter 1170 that includes digital-to-analog converter (D/A), and then to RF using a frequency up-converter. The RF signal is made available at each element using an Analog Distribution Network 1140. Actual transmit beamforming occurs in the far field, as the signal propagates away from the array antenna.

It is well understood that receive array antennas with digitally controlled analog beamformers do not scale well to support multiple beams, as they typically require-one complete analog beamformer including a multi-signal summation network per receive beam and a signal distribution network per transmit beam. They also do not scale well to higher frequencies, as they require dedicated phase shifter hardware including control interface and calibration support electronics, and dedicated low noise amplifier and power amplifier per element.

Antenna Array with Digital Beamformer

FIG. 2 illustrates a prior-art exemplary antenna receive array 200 using a conventional digital beamformer 290. It includes an antenna array 210 with an arbitrary number M of elemental antennas 211-219, each with a dedicated LNA 221-229, and a conventional digital beamformer system composed of dedicated transceivers 231-239 per elemental antenna, each transceiver including a down-converter, an analog-to-digital (A/D) converter, and a digital receiver including memory and an interface to a high-speed digital bus 240. The resulting samples received from each transceiver are stored in the Digital Beamformer Processor 280 and, using the beamforming weights generated by the control processor 250, are used to generate the beams 291-299. It is common knowledge that array antennas with conventional digital beamformers-do not scale well to higher frequencies as they require one full digital receiver, including LNA, a down-converter and analog-to-digital converter per element or sub-array, and extensive support for calibrating and synchronizing such receivers.

FIG. 12 illustrates a prior-art exemplary transmit antenna 1200 array using a conventional digital beamformer 1290 for single or multiple beams. It includes an antenna array 1210 with an arbitrary number M of elemental antennas 1211-1219, each with a dedicated Power Amplifier (PA) 1221-1229, and a conventional digital beamformer system composed of a dedicated digital transmitter 1231-1239 per elemental antenna, each digital transmitter including memory and an interface to a high-speed digital bus 1240, an digital-to-analog (D/A) converter and a frequency up-converter. The Digital Beamformer Processor 1280 uses the beamforming weights generated by the control processor 1250 to create the discrete-time signal composed of the baseband samples for each element, each baseband signal including a combination of data 1291-1299 and weights for the multiple antenna beams.

It is common knowledge that array antennas with conventional digital beamformers do not scale well to higher frequencies as they require one full digital transceiver (i.e., transmitter and receiver), including dedicated LNA and PA, up and down frequency converters, and analog-to-digital and digital-to-analog converters per element or sub-array, and extensive support for calibrating and synchronizing each transceiver.

Conventional phased arrays using element-by-element weighting (amplitude, phase/delay) or digital beamforming (DBF) with a dedicated transceiver (including LNA/PA, up/down frequency converters and A/D and D/A converters) per element do not scale down well in power, real estate area for the electronics, heat dissipation and overall antenna thickness. These conventional antenna solutions do not scale well either to support wideband signals as they worsen the requirements for signal distribution, synchronization, and frequency-conversion, and A/D and D/A speeds.

OBJECTS AND SUMMARY OF THE INVENTION

The objects and summary of this invention will be discussed separately for the receive portion and the transmission portion.

The overall object of this invention is an array antenna with digital beamforming capabilities that uses a single transceiver with one low noise amplifier (LNA)/one power amplifier (PA), one up/down frequency converter and one pair of analog-to-digital and digital-to-analog converters for the entire array. The spread spectrum beamformer method performs the multiplexing of signals to/from each array element with minimum hardware at the array elements and enables the implementation of array antennas with digital beamforming capabilities that are thin and scales large arrays and to high frequencies.

Accordingly, the method object of this invention applies to radars, sensors and communication systems equipped with array antennas. One aspect that differentiates this invention from prior art as applied to digital beamforming is the method of using direct-sequence spread spectrum for both transmission and multiplexing inside arrays with spaced apart array elements. This enables digital beamforming with capability and performance comparable to or better than array antennas using prior art digital beamforming technologies with the advantage that, contrary to prior art, the array antennas DO NOT require a digital transceiver including LNA/PA, up/down frequency converter and analog-to-digital/digital-to-analog converters at each array element or sub-array and, as a result, are simpler to design, manufacture and test; scale up in frequency and in number of elements; scale down in power, weight, thickness and heat dissipation.

One aspect that differentiates the invention from prior art using digitally controlled analog beamforming methods is that it DOES NOT require the use of a low noise amplifier or power amplifier, phase or delay devices, calibration support and control interface per element or sub-array. Another aspect that differentiates the transmit array apparatus and methods object of this invention from prior art using transmit digital beamforming methods, including methods that use time division multiplexing to distribute data and beamforming amplitude and phase weights to the array elements or sub-array, is that it DOES NOT require a dedicated digital transmitter including power amplifier, up-converter, digital-to-analog converter, memory devices, calibration support, synchronization support and high-speed digital interface per element or sub-array. Another aspect that differentiates receive array apparatus and methods object of this invention from prior art using receive digital beamforming methods, including methods that uses time division multiplexing to collect data from the array elements, is that is DOES NOT require a dedicated digital receiver including low noise amplifier, down-converter, analog-to-digital converter, memory devices, calibration support, synchronization support and high-speed digital interface per element or sub-array.

In general, in one aspect included in the preferred embodiment, the required devices per element can be mounted directly at or immediately behind each antenna element, including a patch. This invention includes different spread spectrum methods and array antenna element apparatus for receive beamforming and transmit beamforming.

Receive Beamforming

It is an object of this invention that, when receiving, incident signals at each element are enabled to be multiplexed directly at the frequencies in which they are received, and then extracted without mutual interference while being subject to negligible noise-relate performance degradation. This enables the implementation of low-profile receive array antennas as the low noise amplifier and the digital receiver electronics including down-converter and analog-to-digital converter can be placed remotely from the actual antenna array elements.

Another object of this invention is the spread spectrum receive beamforming method comprising the steps of receiving direct-sequence spread spectrum signals from multiple positions including from a multiplicity of transmitters while using a receive array apparatus composed of spaced elements; reflecting such received spread spectrum signals at each element; combining such reflected signals forming an aggregated signal for the array; converting such an aggregate signal to a convenient intermediary frequency including baseband; sampling such a frequency-converted signal generating a spread spectrum discrete-time signal suitable for digital signal processing; correlating such a spread spectrum discrete-time signal against pseudo-noise sequences or codes derived from the pseudo-noise sequences or codes included in the reflected signals, generating a time-domain discrete-time signal with spread spectrum processing gain that includes both angular and temporal information suitable for spatial demultiplexing; receive digital beamforming including the use of such angular information or corresponding angle-of-arrival hypothesis while achieving array gain including spatial isolation among incident signals; and signal detection using included temporal information while achieving time-and-frequency ambiguity resolution comparable or better than prior-art receive arrays using prior-art analog or digital beamforming techniques, and additional performance gain from multipath or multiple-input/multiple-output (MIMO) information included in the received signals.

Transmit Beamforming

It is an object of this invention that when transmitting, data signals, including amplitude and phase weights for each transmit element or sub-array, are enabled to be multiplexed directly at the frequencies in which they are transmitted. This enables the implementation of low-profile transmit array antennas as the power amplifier and the digital transmitter electronics including up-converter and digital-to-analog converter can be placed remotely from the actual antenna array elements.

Another object of this invention is the spread spectrum transmit beamforming method comprising the steps of generating spread spectrum codes composed of a plurality of pseudo-noise sequences such that each code is a circularly shifted version of every other one of the codes; assigning codes to different array elements, transmit digital beamforming including generating the amplitude and phase ‘weights’ for each antenna element, spreading of resulting discrete-time signals using the assigned codes including equalization for relative synchronization, signal conditioning and weights from additional MIMO encoding; generating an aggregate spread spectrum baseband signal including data, weights and associated codes; up-converting the frequency of such an aggregate spread spectrum baseband signal; power-amplifying such an aggregate RF signal, distributing such an RF signal to the array elements; generating codes at each elemental antenna including a correlating code unique for each antenna and a spread spectrum transmission code; extracting the RF signal for each antenna element including beamformer weights for the element using the correlating code; transmitting the extracted RF signal including beamformer weights while achieving antenna array gain; alternatively spreading and transmitting the extracted RF signal using a common spread spectrum transmission code for the array while achieving transmit array gain; alternatively spreading and transmitting the extracted RF signal using different transmission codes per element while achieving transmit array gain while enabling signal reception methods that include achieving additional performance gain that result from multiple-input/multiple-output (MIMO) processing.

Another object of this invention is the pulse generation and amplification method comprising the steps of coherently amplifying the amplitude of the transmitted signal by coherently combining consecutive segments of the signal extracted at each element including corresponding beamforming weight while using passive components while using a short delay line to store the signal energy contained in such signal segments; generating a sequence of high-power wide-bandwidth short pulses by releasing the signal stored in the delay line over a time period corresponding to the propagation time over the delay line; controlling the amplification of each transmitted pulse by selecting the time interval between consecutive pulses; controlling the short-duration of each transmitted pulse by selecting different lengths for the delay line; transmitting such a sequence of high-power short-duration pulses including multiple short-pulses per chip of resulting spread spectrum signals while achieving transmit array gain and further spreading the bandwidth of the original signal distributed to or generated at each element of the array.

In summary, the invention comprises a receive system and a transmit systems.

The receive system includes a receive array apparatus integrated with a method for generating direct-sequence spread spectrum signals comprised of a multitude of pseudo-noise sequences with a common base sequence including maximal-length sequences with different cyclic shifts, and a method for receiving such spread spectrum signals from different positions including a multiplicity of transmitters at different locations while subjected to reflections from stationary obstacles and moving objects; generating reflected signals at the same carrier frequency as the incident signal while using reflecting signals with distinguishable spectrum features at each array element; combining such reflected signals into a common aggregate radio-frequency signal for the array using either wired or wireless means; converting the aggregate radio-frequency signal to a convenient intermediate frequency including baseband while using a single low noise amplifier and down-converter for the entire array; sampling the resulting aggregate baseband signal while using a single analog-to-digital converter operating at least at the Nyquist rate of the received signals; correlating such an aggregated baseband signal against sequences or variations thereof included in the aggregate radio-frequency signal while capturing angle-of-arrival and multipath information and achieving spread spectrum processing gain and multipath gain; and performing joint beamform and detection to estimate signals, data or parameters included in the incident signals including spatial nulling of a subset of such incident signals while achieving receive array gain. The resulting receive system apparatus and methods enable receive digital beamforming with capability and performance comparable to or better than receive arrays using prior art digital beamforming technologies with the advantage that, contrary to prior art, they DO NOT require a digital transceiver including LNA, down-converter and analog-to-digital converter at each element and, as a result, are simpler to design, manufacture and test; scale up in frequency and number of elements; scale down in power, weight, thickness and heat dissipation.

The transmit system includes a transmit array apparatus integrated with a spread spectrum multiplexing method for generating, aggregating, up-converting, amplifying, distributing and extracting a beamforming radio-frequency signal for each element of the array, a method for re-modulating the radio-frequency signal extracted at each element including corresponding beamforming weight as a direct-sequence spread spectrum signal while using codes that can be equal or different for each element, including codes comprising a pseudo-noise sequence with a different cyclic shift from a common base sequence including a maximal-length sequence, and a method for generating, amplifying and transmitting such extracted radio-frequency signals including resulting re-modulated spread spectrum signals as a sequence of short-duration high-power pulses while using a power amplification method comprised of passive components that combines segmenting a continuous radio-frequency signal in equal-duration wave-blocks, coherently accumulating the energy of each wave-block in a delay line, and then releasing the coherently accumulated energy as short-duration pulse per wave-block while achieving array gain and further bandwidth expansion as compared to the bandwidth of the original signal distributed to or generated at each element of the array. The resulting transmit system apparatus and methods enable transmit digital beamforming with capability and performance including radar-related time-and-frequency ambiguity resolution performance comparable to or better than transmit arrays using prior art digital beamforming technologies with the advantage that, contrary to prior art, they DO NOT require a digital transceiver including power amplifier, up-converter and digital-to-analog converter at each element and, as a result, are simpler to design, manufacture and test; scale up in frequency and number of elements; scale down in power, weight, thickness and heat dissipation.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 shows a prior art example implementation of a receive array antenna apparatus using prior-art digitally controlled analog beamforming technology that requires a low noise amplifier and a phase shifter with associated control interface logic per receive element of the array.

FIG. 2 shows a prior art example implementation of a receive array antenna apparatus using prior-art conventional digital beamforming technology which requires, in addition to a low noise amplifier, a full digital receiver per receive element, including down-converter, analog-to-digital converter and a digital receiver with an interface to a high-speed digital bus and associated synchronization and control interface logic per receive element of the array.

FIG. 3 shows an example implementation of a receive array antenna system using the spread spectrum digital beamforming technology object of this invention in which the signals collected at each antenna element are ‘reflected’ by a pseudo-noise sequence unique to each element, multiplexed using a wired combiner, and processed using a single digital receiver for the entire array including a single low noise amplifier, down converter, analog to digital converter, correlator and beamformer.

FIG. 4 shows an alternative example implementation of a receive array antenna system using the spread spectrum digital beamforming technology object of this invention in which the signals collected at each antenna element are ‘reflected’ through backscattering using a load which impedance is modulated by a pseudo-noise sequence unique to each element, combined wirelessly as the signal propagates towards the digital receiver, and processed using a single digital receiver for the entire array, such a digital receiver including a single low noise amplifier, down converter, analog to digital converter, correlator and beamformer.

FIG. 5 shows an alternative example implementation of a receive array antenna system using the spread spectrum digital beamforming technology object of this invention in which the signals collected at each antenna element are ‘reflected’ through backscattering using a surface acoustic device (SAW) with reflectors that implement a pseudo-noise sequence unique to each element, combined wirelessly as the signal propagates towards the digital receiver, and processed using a single digital receiver for the entire array, such a digital receiver including a single low noise amplifier, down converter, analog to digital converter, correlator and beamformer.

FIG. 6 shows an example implementation of a receive array antenna apparatus using the spread spectrum digital beamforming technology object of this invention in which the signals collected at each antenna element are ‘reflected’ by a pseudo-noise sequence unique to each element, multiplexed and processed by using a single digital receiver for the entire array, such a digital receiver including a single low noise amplifier, down converter and analog to digital converter.

FIG. 7 shows an example of a spread spectrum reflecting code matrix that provides the shift-phase of the reflected spread spectrum signal given for every shift-phase of the reflecting spread spectrum sequence used at the array element.

FIG. 8 shows exemplary spread spectrum code reflections at two exemplary antennas for an incident spread spectrum signal received synchronized with the reflecting codes.

FIG. 9 shows exemplary spread spectrum code reflections at two exemplary antennas when the incident signal is received without chip-level synchronization with the reflecting codes.

FIG. 10 shows an exemplary spread spectrum code reflections at two exemplary antennas for an incident signal containing multiple data element (i.e., a multi-code encoded spread spectrum signal) when the incident signal is received synchronized with the reflecting codes.

FIG. 11 shows a prior art example implementation of a transmit array antenna apparatus using prior-art digitally controlled analog beamforming technology that requires a power amplifier and phase shifter with associated control interface logic per transmit element of the array.

FIG. 12 shows a prior art example implementation of a transmit array antenna apparatus using prior-art conventional digital beamforming technology which requires, in addition to a power amplifier, a full digital transmitter per transmit element, including power amplifier, up-converter, digital-to-analog converter and a digital transmitter with an interface to a high-speed digital bus and associated synchronization and control interface logic per transmit element of the array.

FIG. 13 shows an example implementation of a transmit array antenna system using the spread spectrum digital beamforming technology object of this invention in which the data signal for each transmit element or sub-array, including corresponding amplitude and phase weight, are amplified and multiplexed directly at the frequencies in which they are transmitted, extracted without mutual interference at each transmit element, and then transmitted using its original multiplexing and modulation characteristics including amplitude and frequency modulation, frequency hopping and OFDM multiplexing.

FIG. 14 shows an example implementation of a transmit array antenna system using the spread spectrum digital beamforming technology object of this invention in which the data signal for each transmit element or sub-array, including corresponding beamforming amplitude and phase weight, are amplified and multiplexed directly at the frequencies in which they are transmitted, extracted without mutual interference at each transmit element, and then transmitted as a spread spectrum signal using a common pseudo-noise sequence for the array while maintaining its original amplitude and frequency modulation characteristics.

FIG. 15 shows an example implementation of a transmit array antenna system using the spread spectrum digital beamforming technology object of this invention in which the data signal for each transmit element or sub-array, including corresponding beamforming amplitude and phase weights and additional MIMO weights, are amplified and multiplexed directly at the frequencies in which they are transmitted, extracted without mutual interference at each transmit element, and then transmitted as a spread spectrum signal using different pseudo-noise sequence for each element while maintaining its original frequency modulation characteristics.

FIG. 16 shows an example implementation of the transmitting apparatus at each transmit element in which the extracted signal at each element, including corresponding amplitude and phase weights is transmitted as a sequence of high-power narrow pulses.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

The objects of the invention include a receive array apparatus and receive beamforming methods, and a transmit array apparatus and transmit beamforming methods. FIGS. 3 through 10 apply to the receive system using spread spectrum. FIGS. 13 through 16 apply to the transmit system using spread spectrum multiplexing inside the array. FIGS. 1-2 and 11-12 illustrate receive arrays and transmit arrays using prior-art analog and digital beamforming methods.

Receive Array System

In general, the receive array object of this invention includes a wireless or a wired means for combining the signals reflected at each element while generating a single multi-level direct-sequence spread spectrum aggregate signal at the carrier frequency for the array.

FIG. 3 illustrates a receive array with a wired signal combiner in which the reflected signals result from the multiplication of the reflecting signal at each element with the incident signal. FIG. 4 illustrates a receive array with an alternative wireless signal combiner in which the reflected signals result from backscattering of the incident signal at each element and in which the reflecting signal at each element is used to control the impedance or load at such an element. FIG. 5 illustrates a receive array using the wireless signal combiner in which the reflected signals result from backscattering using SAW devices at each element, each SAW device with acoustic wave reflectors that implement the reflecting signal for such element. FIG. 6 illustrates the receive array including a block diagram of the code reflectors. FIG. 7 illustrates the cyclic shift phase of codes that result from the multiplication (i.e., reflection) of an incident code signal by a reflecting code. FIG. 8 illustrates an exemplary reflected signal that result when the incident code signal is chip-level synchronized with the reflecting codes. FIG. 9 illustrates the same reflection as in FIG. 8 when at least one of the incident signals is not chip-level synchronized with the reflecting codes. FIG. 10 illustrates the resulting reflected signals when the incident signal is itself composed of a multitude of code signals, each code signal with a different cyclic shift phase of a common base sequence.

Receive Array With A Wired Signal Combiner

FIG. 3 depicts the receive array which includes exemplary array antenna 310 with ‘M’ elements and exemplary elements 311-313. The exemplary incident signal 300 includes a direct-sequence spread spectrum signal p_(k)(t) received with cyclic shift-phase ‘k’ relative to the common epoch of the pseudo-noise sequences included in the reflecting signals including exemplary pseudo-noise reflecting sequences 331-333 with exemplary cyclic shift-phases (m−1) through (m+1), each cyclic shift-phase corresponding to one chip delay with duration T_(C) seconds. In the preferred embodiment, the pseudo-noise sequences included in the incident and reflecting signals result from different cyclic shifts of a common maximal-length sequence used as base. The exemplary reflected signals 341-343 result from the multiplication of the incident signal 300 as received at each element with the corresponding reflecting signals 331-333. It is of common knowledge that the multiplication of two pseudo-noise sequences p_(k) and p_(m) from the same family of maximal-length sequences with exemplary—and different—cyclic shift-phases ‘k’ and ‘m’ relative to a base sequence p₀ result in a third maximal-length sequence p_(k@m) from the same family having a third different shift-phase ‘k@m’ where the symbol ‘@’ represents a reflecting cyclic shift-phase operation. It is apparent that the reflected signal at each element #m contain the amplitude ‘γ’ included in the incident signal and the relative phase factor α_(m)=exp[j 2πd/λ sin [θ]] 302 for element #m, in which ‘θ’ is the incident angle, λ is the carrier wavelength and d is the distance between elements. The aggregate signal 349 results from the addition of such reflected signals while using a wired signal combiner using a cable or preferably a microwave signal combiner such as a Wilkinson combiner. In general, the aggregate include additional phase, delay and amplitude variation that result from the interface with and propagation of each component signal in the combiner. In the preferred embodiment, the aggregate signal is received by a single digital receiver composed of a single low noise amplifier 350, a single down-converter 360, and a single analog-to-digital converter 370 operating at the Nyquist rate ‘1/T_(C)’ or higher of the exemplary aggregate signal. The output of the down-converter 361 is at baseband and corresponds to the complex envelope of the aggregate signal. The output of the analog-to-digital converter 371 is the discrete-time version of such baseband signal and includes samples of both the in-phase and quadrature-phase components of the aggregate signal. In general, the down-converter can be implemented in stages including a last stage implemented in software including sampling and analog-to-digital conversion implemented at a convenient intermediary frequency. In the correlator 380, this discrete time baseband signal is correlated against all pseudo-noise sequences included in the reflected signals including corresponding base sequence with every possible cyclic shift. It is of common knowledge that maximal-length sequences ARE NOT truly shift orthogonal as they have a non-zero off-phase cyclic auto-correlation. In the preferred embodiment using maximal-length sequences, such an orthogonalization is achieved by using a variation of the maximal length sequences when performing the correlation operations, such a variation resulting from using for each cyclic shift just the chip positions corresponding to positive values of the resulting shifted maximal-length sequence. In the exemplary case of FIG. 3 in which the incident signal includes a single pseudo-noise sequence with shift-phase ‘k’, the reflected signal from an element with reflecting signal using shift-phase ‘m’ results with a shift-phase ‘k@m’. In the case illustrated in FIG. 3 in which the chips of the incident signal 300 are synchronized with the chips of the reflecting signals 331-333, there is a one-to-one correspondence between each antenna element 311-313 and a corresponding detected signal position at the output of the correlator. Specifically, the correlator output 381 corresponding to element number ‘m’ for an incident signal with shift-phase ‘k’ results at position ‘k@m’ with spread spectrum processing gain (L+1)/2 while having a complex-valued amplitude that include the complex-valued amplitude ‘γ’ of the incident signal and the phase factor α_(m) for the element number ‘m’. In this case, in which the incident signal is composed of a single pseudo-noise sequence and the resulting correlation operations are orthogonal for different cyclic shifts, the beamformer 390 uses conventional beamforming methods and weights to generate the beamformed signal 391 with array gain ‘M’ including placing antenna nulls (i.e., null steering) as desired, including nulls toward selected incident signals including signals from jammers.

Receive Array with a Wireless Signal Combiner

FIG. 4 depicts the receive array 400 using a wireless signal combiner in which exemplary reflected signals 441-443 are generated using a switched-load modulation method controlled by the reflecting signals 431-433 in which a load with an impedance including short and open selected such to maximizes signal reflection through backscattering. In the preferred implementation, using binary reflecting signals, including reflecting signals derived from a base sequence that is a maximal-length sequence with chip values ‘±1’, such a differential delay is set to 180° corresponding to one half of the carrier wavelength. As it well understood, the reflected signals propagate through backscattering from the elements to a common receiver 444 located somewhere else generating the aggregate reflected signal 449, such a receiver including the receiving devices 450-490 and using the amplification, down-converting, correlating and beamforming methods as described in conjunction with FIG. 3

FIG. 5 depicts the receive array 510 which also uses a wireless combiner, the exemplary reflected signals 541-543 for an incident signal 500 are generated using surface acoustic wave (SAW) devices, 521-523, which include an inter-digital transducer (IDT) 534 which converts electromagnetic waves into acoustic waves using piezoelectric effects and a set of wave reflectors 531-533. The configuration of these wave reflectors varies from element to element such as to produce reflecting structures that correspond to the exemplary reflecting codes ‘p_((m−1))’ through ‘p_((m+1))’ and generate reflected signals with characteristics as described in conjunction with FIGS. 3 and 4. As in the system of FIG. 4, these reflected signals propagate through backscattering 540 from the elements to a common receiver 554 located somewhere else while producing the aggregate reflected signal 549, such a receiver including the receiving devices 550-590 and using the receiving and processing methods as described in conjunction with FIGS. 3 and 4.

Preferred Embodiment Using a Wired Signal Combiner

FIG. 6 illustrates an exemplary antenna array apparatus 610 using the preferred embodiment for the spread spectrum beamformer method object of the invention. In general, the incident signal 600 includes a multitude of maximal-length sequences from a common base sequence with different cyclic shifts. The receive array apparatus includes an antenna array 610 with an arbitrary number M of element antennas 611-618, each with dedicated code reflectors 620, each code reflector 621-628 including a shift register ‘PN GEN 1’ through ‘PN GEN 8’ used to generate the a maximal length reflecting sequence 631-638, a device 639 such as a mixer or quadrature hybrid such as a Lange coupler used in conjunction with a switched load that includes ‘short’ and ‘open’ to multiply including modulate the received signal by the generated maximal-length sequence, and an interface to an analog signal combiner 640 such as a Wilkinson combiner used to aggregate the reflected signals 641-648 reflected at each element. In the preferred embodiment, the aggregate signal 649 is received by a single digital receiver composed of a single low noise amplifier 650, a single down-converter 660, and a single analog-to-digital converter 670 operating at the exemplary Nyquist rate (1/Tc) or higher of such an aggregate signal. The output of the down-converter 661 is at baseband and corresponds to the complex envelope of the aggregate signal 649. The output of the analog-to-digital converter 671 is the discrete-time version of such baseband signal and includes samples of both the in-phase and quadrature-phase components of such an aggregate signal. This discrete time signal is correlated against the corresponding base sequence including every possible cyclic shift. The detected signals 681-688 at the output of the correlator 680 result with spread spectrum processing gain and includes information of the data included in the transmitted signals as well as information of the relative phase and amplitude in which each the incident signal 600 is received at each element 611-618. These relative phase and amplitude are used by the Beamformer Processor 690 to jointly perform the spatial demultiplexing of a multitude of signals incident at the array from different angular directions or antenna beams 691-192, including performing joint detection and beamforming of multiple data included in each incident signal included in each antenna beam.

Reflected Codes with Maximal-Length Sequences

FIG. 7 illustrates the cyclic shift-phase relationship between incident, reflecting and reflected signals for an exemplary maximal-length sequence with length 15 in which each row corresponds to a code with exemplary cyclic shift-phase 711 of the incident signal, each column corresponds to a code with exemplary cyclic shift-phase 733 of the reflecting signal at an exemplary element, and each row-column intersection corresponds to the code with exemplary cyclic shift-phase 753 of the corresponding reflected signal.

In general, in one aspect relevant for the orthogonality of the correlation operations, the codes or relative cyclic shift-phases of pseudo-noise sequences included in the reflecting signals are selected such as to be different from the codes or relative shift-phases included in the incident signals, such shift-phases or codes being defined relative to a referential time-epoch in which the samples corresponding to such incident pseudo-sequences are first sampled and then selected for processing using the correlation operations.

Spread Spectrum Multiplexing with Synchronization

FIG. 8 illustrates an exemplary direct-sequence spread spectrum pseudo-noise sequence incident signal 800 being reflected at two exemplary elements #1 and #2 while using exemplary reflecting signals 831 and 832 with cyclic shift-phases 831A and 832A respectively, in which such cyclic shift-phases are defined relative to a referential time or code epoch 871. In the example, the cyclic shift-phase 801 of the incident signal relative to such referential time-epoch is ‘2’; the cyclic shift-phase 831A of the reflecting signal 831 for element #1 is ‘0’, and the cyclic shift-phase 841A of the resulting reflected signal 841, corresponding to the value in the intersection of row#3 and column #1 in the table of FIG. 7, is ‘9’. Similarly, the cyclic shift-phase 832A of the reflecting signal 832 for element #2 is ‘1’ and the cyclic shift-phase 842A of the corresponding reflected signal 842, corresponding to the value in intersection of row#3 and column #2 in the table of FIG. 7, is ‘13’ (or ‘D’ in hexadecimal).

Spread Spectrum Multiplexing Without Synchronization

In general, the incident signal includes at least one pseudo-noise sequence or code that is not synchronized at the chip-time level with the referential-time epoch of the receive array. FIG. 9 illustrates an exemplary case of such a non-synchronized incident signal 900 composed of a single pseudo-noise sequence including cyclic padding. As in the example of FIG. 8, such an incident signal is reflected by the exemplary reflecting signals 931 and 932 each with cyclic shift-phases 931A (with cyclic shift-phase ‘0’ at element #1) and 932A (with cyclic shift-phase ‘1’ at element #2) respectively. In this case, and as illustrated in the figure, each element generates a pair of reflected signals 941A/B and 942A/B, each with different cyclic shift-phase offsets 951A/B and 952A/B corresponding respectively to cyclic shift-phases ‘9’ and ‘4’ at element #1, and cyclic shift-phases ‘13’ (“D” in hexadecimal) and ‘10’ (‘A’ in hexadecimal) at element #2. As in the example of FIG. 8, the cyclic shift-phase values of the reflected signals are determined by shift-and-add reflecting properties of the maximal-length sequences summarized in the table of FIG. 7. The power of each reflected signal varies a function of two parameters: (1) the fractional-chip time-offset between the pseudo-noise sequences included in the incident signal and the pseudonoise sequence of each reflecting signal; (2) the overall impulse response (not shown) of the transmit-receive channel for each antenna element including filtering effects of the antenna elements themselves, filtering effects added during transmission, and filtering effects of additional active and passive circuitry used between each antenna element and the output of the signal combiner. In general, in one aspect that is relevant for beamforming, both the relative synchronization offset and the relative carrier phases 961 and 962 in which the signal incident with an exemplary angle ‘θ’ with respect to the boresight of the array is received at each element are used to determining the beamforming weights used generate the beamformed signal or signals toward specific incident angular directions, including nulls and/or determining the beamforming/detecting matrix used to jointly beamform and detect data or other information-bearing parameters included in the incident signal 900.

Support for Spread Spectrum Signals with Multiple Codes

The methods of the invention include support for spread spectrum incident signal with multi-code data. FIG. 10 illustrates an exemplary receive array with a signal 1000 incident at an angle ‘θ’ and including two superposed exemplary pseudo-noise sequences with different cyclic shift-phases 1011 and 1012, each used to ‘spread’ different exemplary sensors samples or modulated data symbols a₁ and a₂. Also illustrated are the reflecting signals 1032 and 1036 with cyclic shift-phases 1032A and 1036A respectively at two exemplary elements ‘#2’ and ‘#6’. As in the example of FIG. 8 using maximal-length sequences, each of such incident pseudo-noise sequence generates at each element a reflected signal with cyclic shift-phase as specified in FIG. 7. Specifically, at element #2 using a reflecting signal 1032 with shift-phase 1032A (actually ‘1’), the incident pseudo-noise sequence 1000A with cyclic shift-phase 1011 (actually ‘2’) produces a reflected signal 1042A with cyclic shift-phase 1052A (actually ‘13’ or ‘D’ in hexadecimal) while the other pseudo-noise sequence 1010B with shift-phase 1012 (actually ‘3’) produces a reflected signal 1042B with shift-phase 1052B (actually ‘10’ or ‘A’ in hexadecimal). The resulting reflected signals at element #2 include a phase factor in the form of a complex-valued amplitude 1062 that captures the relative phase in which the carrier is received at element #2 relative to element #1 (not shown). Similarly, for element #6, using a reflecting signal 1036 with shift-phase 1036A (actually ‘5’), the incident pseudo-noise sequence 1000A with shift-phase 1011 (actually ‘2’) produces a reflected signal 1046A with shift-phase 1056A (actually ‘6’) while the other pseudo-noise sequence 1000B with shift-phase 1012 (actually ‘3’) produces a reflected signal 1046B with shift-phase 1056B (actually ‘12’ or ‘C’ in hexadecimal). The resulting reflected signals at element #6 include a phase factor in the form of a complex-valued amplitude 1066 that captures the relative phase in which the carrier is received at element #6.

In general, the set of reflected signals at different elements resulting from incident signals with code division multiplexing including multiple superposed pseudo-noise sequences with different cyclic shift-phases, will result with different spatial-temporal signatures in which the spatial-temporal signature for each element includes the set of cyclic shift-phases included in the corresponding reflected signals and the complex-valued amplitude including relative carrier phase in which each multi-code incident signal is received at such an element. Using prior-art technologies including prior-art of multiple-input/multiple-output (MIMO) systems, it is well understood that such a set of different spatial-temporal signatures can be used for the demultiplexing of incident signals resulting from multiple transmitters and alternatively (or jointly), for detecting multiple modulated data included in the incident signals resulting from each transmitter. It is also well understood in such prior art technologies that the mutual correlation among the spatial-temporal signatures can greatly impact the performance of such demultiplexing and detection operations. In the preferred embodiment, the reflecting sequences at each antenna element are multiplied by a fixed but otherwise random sequence of constants including the values ±1 such as to maximize the signal-to-noise-plus-interference ratio (SNIR) of the signal detection methods, including methods for detecting signals with multiple-input/multiple-output (MIMO) characteristics.

Transmit Array System

In general, the transmit array object of this invention includes means for multiplexing the data and amplitude and phase weights for each element as a multi-level aggregate signal at a carrier frequency including frequency modulation or frequency hopping, means for distributing the signals to each transmit element, means for extracting such data and amplitude and phase weight at each element without mutual interference, means for alternatively re-encoding the resulting extracted data signal as a direct-sequence spread spectrum signal, means for compressing the chips of such spread spectrum signal as time-compressed pulses with higher bandwidth than the original spread spectrum signal and high instantaneous power than the chips of such spread spectrum signal, and means for transmitting such resulting extracted data signal including additional re-encoding and time-compressed pulses while achieving array gain, spread spectrum processing gain and addition time-compressing performance gains, including SNR gain and, for radar applications, ambiguity resolution gain.

FIG. 13 illustrates a transmit array with a wired signal combiner in which the signal extracted and transmitted by each element signals result from the multiplication including demodulation or re-modulation of the spread spectrum multiplexed aggregate signal with a code generated at each element, and that is then transmitted as extracted by each element. In this case, the aggregate direct-sequence spread-spectrum multiplexed signal sent to the various elements, in addition to the beamforming weight for each element, may include additional amplitude modulation, frequency modulation or frequency hopping. FIG. 14 illustrates a transmit array in which the signals extracted at each element, are re-encoded as a spread spectrum signal using a code common for the entire array. FIG. 15 illustrates a transmit array that can be used in radars and communications systems with multiple-input/multiple-output (MIMO) characteristics in which the signals sent and extracted at each element have different amplitudes, and the signal transmitted by such elements have different spread spectrum codes. FIG. 16 illustrates a method to compress and transmit the extracted signal at each element including chips of resulting spread spectrum signals as a sequence of narrow high-power pulses.

In transmit system of FIG. 13, the aggregate baseband signal 1331 for the array includes the modulating signal ‘a(t,w)’ and the beamforming weights {α*₁, α*₂, . . . } 1320 multiplexed using the exemplary codes ‘p_(k)(t)’ 1310, including codes with cyclic shift-phase ‘k’ relative to a common code epoch, each cyclic shift-phase corresponding to one chip delay with duration ‘T_(C)’ seconds. In the preferred embodiment, the pseudo-noise sequences result from different cyclic shifts of a common maximal-length sequence used as base sequence. The aggregate baseband signal including synchronization delay τ_(k) for each element is amplified and up-converted to the carrier frequency ‘w’ included in the up-converted aggregate signal 1361 using a common-for-the-array Digital-to-Analog converter 1340, a common-for-the-array up-converter 1350, and a common-for-the-array power amplifier 1360. The up-converted aggregate signal is distributed to the various elements using a distribution apparatus 1370 that includes power dividers, cascade in-series signal distribution and, for a reflecting array (not shown), over-the-air transmission from a common feed-point including an antenna horn. At each element an exemplary pseudo-noise code generator 1380 is used to generate the exemplary correlating codes 1381-1383 derived from the base sequence and included in the aggregate signal. The demodulation or de-spreading of the signal at each element is performed using a multiplication including the code reflectors of FIG. 6 followed by a band-pass filter 1390. The codes and the synchronization delays τ_(k) are selected such as the de-spread signals at the output of the band-pass filter 1390 result, with no interference from signals for the other elements and included in the aggregate signal. Ideally, the signal extracted at each element results as a continuous carrier at a common frequency ‘w,’ includes the amplified version ‘A(t,w)’ of the baseband signal component ‘a(t,w)’ common for the entire array and the beamforming weights ‘α*₀(t,w)-α*_(M−1)(t,w)’ 1391-1393 and, as shown in 1394, specific for each array element such that the beamformed signal result with array gain ‘M’ 1399 at the angular direction ‘θ.’ 1398.

FIG. 14 illustrates a variation of the transmit array of FIG. 13 in which the resulting signals 1491-1493 at each array element result from the multiplication or re-modulation of the signal at the output of the band-pass filter 1490 are spread by a pseudo-noise sequence ‘q(t)’ 1484 that, although generated locally at each element, is common for the whole array, as shown in 1491-1493. In the preferred embodiment, the demodulation or de-spreading of the aggregate signal 1461 at each array element using the codes 1481-1483, and re-modulation or re-spreading of the signals at the output of the bandpass filter by the common code 1484 common for the array include the code reflecting methods of FIG. 6. Also, the code 1484 common for the whole array can be equal or not to one of the codes included in the baseband signal 1431.

As it well understood, the beamformed signal 1496 at the direction ‘θ’ 1498, in addition to resulting array gain ‘M’ 1499 proportional to the number of elements in the array, retains the instantaneous carrier frequency ‘w’ and instantaneous amplitude A(t,w) of the aggregate signal 1461.

FIG. 15 illustrates yet another variation of the transmit array of FIGS. 13-14 in which the amplitudes ‘A_(k)(t,w)’ 1561 and the codes ‘q_(k)(t)’ 1584-1586 included in the signals 1591-1593 transmitted by each element are different from element to element. In this case, in the preferred embodiment, when used in conjunction with multiple-input/multiple-output (MIMO methods, the codes 1584-1586 are drawn from codes included in FIGS. 3-10 including codes derived from maximal length sequences including methods for detecting such codes without mutual interference.

FIG. 16 illustrates a variation of the circuitry for a generic array element number ‘m’ of FIGS. 13-15 that is relevant for wideband applications including radar and communications in which the signals extracted at each element are transmitted as a sequence of high-amplitude pulses with short-duration and consequent higher bandwidth than the original signals extracted at each element. The aggregate up-converted signal 1661, after being multiplied or re-modulated by code 1681 using the multiplication circuitry 1682 that includes the code reflecting method of FIG. 6, and then passed through the band-pass filter 1690 generates the extracted component signal 1691 for the element that includes mainly a signal at the instantaneous carrier frequency ‘w’ common for the array that includes the amplitude A_(m)(t,w) and the phase factor α*_(m)(w) for the element. This signal, including possible additional re-modulation by code ‘q_(m)(t)’ 1684 as described in conjunction with FIGS. 14 and 15, is the input for the pulse generator 1692. In the preferred embodiment, this pulse generator is composed of a signal combiner 1693, a quadrature hybrid 1694 that includes a Lange coupler, and a pair of delay lines 1695A and 1695B shortened to ground, each with a round-trip delay corresponding to an odd number (2N+1) of quarter-long wavelengths (i.e., λ/4). As it is well understood, the signal at the output port #4 of the quadrature hybrid 1694 can be made to be an exact in-phase replica, although slightly attenuated, of the input signal at the input port #1. The delayed signal 1662B at the output port #4 adds coherently with input signal 1662A such that signal at the output of the combiner 1693 results magnified by a factor β included in the output signal 1698, such a magnifying factor being proportional to the duration of an wave-block T_(B) and the round-trip delay [(2N+1)/4 T_(λ)] in the delay line 1695A or 1695B, where ‘T_(λ)=2π/w’ is the period of the carrier signal at frequency ‘w’. As it is well understood, the signals 1662A and 1662B include the amplitude A_(m)(t,w) and phase factor α_(m)(t,w) that, for the communications and radar applications object of this invention remain constant over the duration T_(B) of a wave-block 1697. At the end of each wave-block, the switch at the output of the combiner 1693 switches from the input to the hybrid 1694 to the output line 1696 that connects directly to the radiating element including a patch 1699. It is well understood that for the exemplary spread spectrum signal and parameters included in FIG. 16, the signal 1698 radiated by the element antenna 1699 is composed of a train of short-duration pulses per chip-time ‘T_(C),’ including a single pulse per such a chip time, each pulse with an amplitude corresponding approximately to the ratio between the duration ‘T_(B)’ of each wave-block and the round-trip propagation in the delay lines 1695A or 1695B. As it is well understood for radar applications, such a radiated signal with short-pulses, each with magnified amplitude, results in radar system with enhanced performance including enhanced time ambiguity resolution.

The following paragraphs summarize features and characteristics of signals, devices or apparatus included in the preferred implementation.

Transmitted Signals

The following applies to signals transmitted by the array antenna using the spread spectrum beamformer object of this invention. In general, the invention includes a spread spectrum beamforming method to multiplex, distributed and extract signals for each antenna element, and means for re-modulating and controlling the temporal and spectral characteristics of signals radiated by transmit array while achieving array gain and beamforming with user defined characteristics including multiple beams and antenna nulls:

-   -   In one aspect included in the preferred embodiment and relevant         for communications, radar and RF sensing applications, the         transmitted signal includes support for frequency modulation         including frequency hopping and OFDM, and support for         direct-sequence spread spectrum including signals with         additional MIMO encoding and signals with shift-orthogonal         characteristics suitable to be received and processed by the         receive arrays using the spread spectrum receive beamforming         method included in this invention.

Support for Multi-Bit Data Communications

-   -   In one aspect included in the preferred embodiment and relevant         for data communications, the transmitted spread spectrum signal         includes code division multiplexing using superposed         pseudo-noise sequences or codes derived from a common base         sequence, each with a different cyclic shift-phase and used to         spread a different modulated symbol, including symbols resulting         from BPSK, M-ary PSK and M-ary QAM modulations.

Support for Sensor Applications

-   -   In one aspect included in the preferred embodiment and relevant         for sensor data dissemination, the transmitted spread spectrum         signal includes code division multiplexing using superposed         pseudo-noise sequences or codes derived from a common base         sequence, each with a different cyclic shift and used to spread         a different complex-valued sample resulting from sample-and-hold         apparatus included in sensor devices.

Support for Wideband Radar Applications

-   -   In one aspect included in the preferred embodiment and relevant         for high-power wideband radar applications, the re-modulated         signals including signals with frequency modulation or frequency         switching including frequency hopping, frequency multiplexing         including OFDM, and direct-sequence spread spectrum including         MIMO encoding are transmitted as a sequence of short-duration         high-power RF pulses including one or multiple RF pulses per         chip, such short-duration RF pulses a few carrier cycles per RF         pulse and generated at and/or immediately before the radiating         element, such high-power generated without active-circuit         amplification through passive circuit means performing coherent         energy accumulation over a wave-block with user or         application-defined length in time including a chip-time of a         spread spectrum signal.

Support for Multi-Function Adaptive Radar Applications

-   -   In one aspect included in the preferred embodiment and relevant         for radar applications including performance adaptation and         optimization for searching, tracking and multiple target         detection, isolation and tracking, the type of waveform         transmitted by the array is user defined including flexible         frequency modulation including frequency hopping and OFDM,         flexible spread spectrum including support for MIMO, and         flexible transmission of such waveforms using coherently         amplified short-duration pulses including pulses with a couple         or few carrier cycles per pulse.

Support for Flexible Beamforming and Encoding

The following applies to the array patterns including power profile including size and localization of sidelobes of the beamformed beams, and types of re-modulation and power-profile of transmitted signals. In general, the array-weights generated and extracted at each element can have any combination of amplitude, phase and delay thus enabling flexible beamforming while the types of transmitted signal can vary from continuous frequency-modulated signals to spread spectrum signals transmitted with short-duration pulses:

-   -   In one aspect included in the preferred embodiment and relevant         for communications and radar applications with flexible         beamforming including generating multiple beams, multiple nulls         and controlling the shape of such beams and power of sidelobes,         the beamforming weights generated for and extracted at each         element, have user-defined amplitudes, phases and delays.     -   In one aspect included in the preferred embodiment and relevant         for communications and radar applications with flexible spread         spectrum encoding including encoding for MIMO processing, the         signals including amplitudes and spread spectrum signals         transmitted by each element can be different for each element.

Low Profile and Scalability in Frequency

The following applies to the localization of the power amplifiers and the active/passive nature of the circuits and components per array element. In general, the power amplification of signals distributed to the various elements is performed removed from the array, and the circuits used to extract, re-modulate, re-shape, and further amplify the signal at each element is performed using passive components:

-   -   In one aspect included in the preferred embodiment and relevant         for array systems with that are thin and operate at high         frequencies, the power amplification of the transmitted signals         is performed physically removed and before the signals are         distributed to the various elements and the processing and/or         signal conditioning at the elements are performed using         RF-passive circuits including signal extraction at each element         performed by re-modulation and filtering using RF-passive         components, encoding for spread spectrum transmission through         re-modulation using RF-passive components, and short-pulse         generation and amplification through coherent carrier         accumulation over a wave-block using RF-passive components;     -   In one aspect included in the preferred embodiment and relevant         for radar and communications applications using high-powered         short-duration pulses, the pulse generator of FIG. 16 includes         the use quadrature hybrids, delay lines, combiner and microwave         circuits implemented using super-conductors, including         room-temperature superconductors, such as to minimize losses and         maximize the amplitude of such transmitted pulses;     -   In one aspect included in the preferred embodiment and relevant         for power-efficient amplification, the transmitted spread         spectrum signal includes two or more component signals resulting         from a phase splitting decomposition, each component signal with         power amplification characteristics including having constant         power envelope suitable for amplification by power amplifiers         operating at or at-near saturation.

Compatibility with Spread Spectrum Receive Beamforming

The spread spectrum signals generated, transmitted and beamformed by the transmit array include a multitude of superposed maximal-length sequences from a common base such as they can be receive and processed by the spread spectrum receive beamformer methods object of this invention.

Received Signals

The following applies to signals that are received by the array antenna with spread spectrum beamformer object of this invention. In general, the signals incident at the receive array include spread spectrum signals from one or multiple transmitters or from multiple targets located at different positions:

-   -   In one aspect included in the preferred embodiment and relevant         for radar, communications and sensor applications, each of such         incident signals may include multipath components resulting of         reflections from field or urban structures such as mountains and         buildings;     -   In one aspect included in the preferred embodiment and relevant         for radar tracking multiple targets at different positions, each         of incident signals may result from a multitude of stationary or         moving object such as airplanes in multiple locations;     -   In one aspect included in the preferred embodiment and relevant         for fine time-and-frequency ambiguity resolution, the spread         spectrum encoding of incident signals may be include chips, each         chip composed of one or a multitude of short duration pulses;

Reflecting Signals

The following applies to the reflecting signals used inside the receive array in conjunction with the spread spectrum beamformer object of this invention. In general, the receive array include spaced elements, each element with an apparatus to reflect the incident signal using a different reflecting signal such that the reflected signals from different elements have mutually distinguishable features:

-   -   In one aspect included in the preferred embodiment and relevant         to minimizing the electronic support at the element, the         reflection method include means for generating reflected signals         with distinguishable complex envelope features including         relative amplitude, delay or phase while maintaining the carrier         frequency and relative carrier phase of each of the incident         signals for at least the duration of one pseudo-noise sequence         included in such incident signals;     -   In one aspect included in the preferred embodiment and relevant         for generating reflected signals with distinguishable features,         the reflecting signal at each element having distinguishable         spectrum features, including different amplitudes, phases or         delays including cyclic-shift delays;     -   In one aspect included in the preferred embodiment and relevant         for subsequent correlation and detection operations, the         reflecting signal at each element use different spread spectrum         signals including spread spectrum signals with different         integer-chip delays, including pseudo-noise sequences of a         common base that repeat cyclically, each with a different cyclic         shift-phase, including pseudo-noise sequences with the same base         as the pseudo-noise sequences included in the incident signals;     -   In one aspect included in the preferred embodiment and relevant         for the orthogonal correlation of different pseudo-noise         sequences included in the reflected signals, the reflecting         signal at each element is a maximal-length sequence with         different cyclic shift-phase that have the same base as the a         maximal-length sequences included in the incident signal, each         having a different cyclic-phase than the cyclic phases included         in the incident signal;     -   In one aspect included in the preferred embodiment and relevant         for the practical realization of the reflecting signals at each         element, the reflecting maximal-length sequence at each element         is implemented using linear feedback shift register (LFSR)         techniques in which the shift-register of each element is         initialized at a different initial state, such an initial state         corresponding to the cyclic shift-phase for such element;     -   In one aspect included in the preferred embodiment and relevant         for extracting the signal incident at each element or for         performing the beamforming operation, the reflecting signal at         each element include overlaying a random factor including         selecting the ±1 sign or 180° phase-shift of the corresponding         reflecting signal including means for programming the overall         initial state of the LFSR at such an element;     -   In one aspect included in the preferred embodiment and relevant         for the simplification of the beamforming operations, the LFSR         of every element included in the receive array is driven from a         common clock signal synchronized with the clock signal used in         the sample-and-hold device included in the analog-to-digital         converter common for the entire array;     -   In one aspect included in the preferred embodiment and relevant         for range measurements, the clock that drives the generation of         the reflecting sequences at each element, and the generation of         the reflecting sequences themselves are synchronized with code         epoch common for the whole array;     -   In one aspect included in the preferred embodiment and relevant         for maximizing the number of elements in the array, the         transitions of the clock signal used to drive the LFSR at each         element is off-phase with the transitions of the clock signal         used in the sample-and-hold device included in the         analog-to-digital converter common for the entire array.     -   In one aspect included in the preferred embodiment and relevant         for high-resolution of time ambiguities and for support         applications using applications short pulses as per descriptions         of methods and procedures corresponding to FIG. 16, the chip         clocks and the synchronization signals have characteristics         compatible with the use of such short-duration pulses;     -   In one aspect included in the preferred embodiment and relevant         using a single LNA for the entire array and for minimizing the         minimizing the hardware requirements at each element including         overall volume, heat dissipation and power consumption, the code         generation and reflecting is implemented using the code         reflector method of FIG. 6 requiring no active RF hardware.

Reflected Signals

The following applies to the apparatus and method used to reflect the signals incident in the receive array object of this invention and with the method and means to combine such reflected signals such they can be collected at common point including the interface with the down-converter. In general, the receive array object of this invention includes a wireless or a wired means for combining the signals reflected at each element while generating a single multi-level aggregate signal at a common carrier frequency for the entire array:

-   -   In one aspect relevant for the wireless combining of the         reflected signals, such a signal reflection includes reflection         through backscattering in which part of the incident signal at         each element is reflected back due an impedance mismatch between         the antenna element and the load circuit in which the reflecting         signal is used to control such a load, including controlling the         switching between different loads, or controlling the switching         between loads subjected to different delays with such delays set         or measured relative to the phase center of the corresponding         antenna element, or controlling the resistance or impedance of         variable resistance devices such as PIN diodes;     -   In another aspect relevant for the wireless combining using         signal reflection through backscattering at each element         includes the use of surface acoustic wave (SAW) devices,         including SAW devices with inter-digital transducer and a set of         wave reflectors that produce a unique sequence of reflected         acoustic wave pulses that, after being converted to a sequence         of radio pulses, including a pseudo noise sequence of such radio         pulses, are transmitted back through the antenna element;     -   In one aspect included in the preferred embodiment and relevant         for using wired signal combiners such as a Wilkinson combiner,         the signal reflection method includes devices that use the         reflecting signal to control the amplitude and phase         characteristics of the incident signal including devices that         perform the multiplication or re-modulation of the incident         signal by the reflecting signal at each element;     -   In one aspect included in the preferred embodiment and relevant         for reflecting signals that are binary such as maximal length         sequences, the method used to multiply or re-modulate the         incident signal by the reflecting signal includes the method         described in conjunction with FIG. 6 that including the use of         quadrature hybrids and reflecting the incident signal by 0° or         180° according to the ±1 values included in the reflecting         signal;     -   In one aspect included in the preferred embodiment and relevant         to reducing the noise of such switching operations in the         reflected signals includes using reflecting signals that are         synchronized with sample-and-hold devices included in the         analog-to-digital converter such a sampling being performed with         a phase offset relative to consecutive switching times,         including half-cycle between such consecutive switching times.

Digital Receiver

The following applies to the apparatus and method used to amplify, down-convert and sample the aggregate radio signal resulting from the combination of reflected signals. In general, such an aggregate signal has the same carrier frequency as the incident signal and a higher bandwidth:

-   -   In one aspect included in the preferred embodiment and relevant         for noise performance and equipment count or cost reduction, the         aggregate radio signal resulting from either the wired or         wireless combination of the reflected signal is fed to and         processed by a single low noise amplifier followed by a single         down-converter, such a down-converter used to create an         intermediate-frequency signal or baseband signal corresponding         to such an aggregate signal;     -   In one aspect included in the preferred embodiment and relevant         for noise performance and equipment count or cost reduction, the         down-converted signal either at baseband or at an intermediate         frequency is, sampled while using a single or a pair of         analog-to-digital converters operating at least at the Nyquist         rate of the aggregated reflected signal, such analog-to-digital         converter devices used to generate a complex-valued (i.e.,         in-phase and quadrature-phase) digitized discrete-time version         of the aggregate signal suitable for processing by a general         purpose computer or digital signal processor devices.

Correlator

The following applies to the correlation operations performed using the baseband samples of the aggregate signal that resulted from the combination of the reflected signals. In general, the signal at the output of the analog-to-digital converter results from the combination of multiple reflected signals, each reflected signal being a direct-sequence spread spectrum signal including at least one signal including at least one pseudo-noise sequence or code:

-   -   In one aspect included in the preferred embodiment, the cyclic         correlation operations are used to transform the resulting         baseband sampled signal from the code-domain (i.e., spread         spectrum) to the time-domain (i.e., time division multiplexing)         while achieving spread spectrum processing gain. In the         preferred embodiment, the signal that result at the output of         the correlator is the discrete-time version of such a         time-domain baseband signal corresponding to the radar waveform,         analog sensor signal or modulated data signal included in the         spread spectrum signal and used to spread the pseudo-noise         sequences included in the incident signal;     -   In one aspect included in the preferred embodiment, the         resulting baseband signal is a code-domain signal that includes         at least one code or cyclic shift of the maximal length sequence         with ±1 chips and used as base sequence included in at least one         of the incident signals;     -   In one aspect included in the preferred implementation and         relevant for the orthogonal detection of the spread spectrum         codes included in the aggregate signal, the resulting baseband         signal is correlated against all cyclic shifts of a modified         version of the maximal length sequence used as base sequence and         included in at least one of the incident signals, such a         modified version of such a maximal-length sequence having each         chip-position with a negative value replaced by a zero.

Signal Detection and Beamforming

The signal at the output of the correlator resulting from the correlating operations is a time-domain signal as it includes the baseband waveform, samples or modulated symbols used to spread the pseudo-noise sequences used to compose the transmitted spread spectrum signal. In certain embodiments, including embodiments in which the incident signals have different codes, such a time-domain signal at the output of the correlator is such that it can be spatially demultiplexed while using information of the angle-of-arrival of each incident signal:

-   -   In one aspect included in the preferred embodiment, the digital         beamforming operations include the use of weights that include         information of one, a set or all the incident angles from         potential transmitters or targets, to separate signals incident         at different angles while achieving array gain proportional to         the number of elements;     -   In one aspect included in the preferred embodiment, the digital         beamforming operations include the use of weights that include         information of one, a set or all the incident angles, to         generate antenna nulls in specific directions, including         directions corresponding to the angles-of-arrival of selected         incident signals.

In general, due propagation channel memory including multipath, or when the transmitted signals include multiple codes, or the incident signals from different transmitters includes at least one pseudo-noise sequence with a common cyclic shift-phase in at least two of such incident signals, the reflected signals by at least two of the elements include at least one pseudo-noise sequence with a common cyclic shift-phase or code:

-   -   In one aspect included in the preferred embodiment and relevant         for performance maximization, the methods object of this         invention include techniques for joint beamforming and detection         include multiple hypothesis tests and inversion techniques         including pseudo-inversion operations leveraging         angle-of-arrival information or hypothesis, multipath         signatures, additional beamforming weights including weights         introduced at the reflecting sequences to maximize the noise         performance of the joint detection methods using         pseudo-inversion techniques.

In general, at least one of the incident signals are not chip-level synchronized with the reflecting signals of at least one element, generating corresponding reflected signals that include at least two codes with different code-shifts as per exemplary FIG. 7, each at least one sequence with different non sequential non-contiguous cyclic shifts, each sequence with different power. These different codes with different cyclic-shifts, after the correlating operations result in non-contiguous pulses (pulse splitting):

-   -   In one aspect included in the preferred embodiment and relevant         for performance maximization, the methods object of this         invention include using the relative position, form and power of         such split pulses to estimate the signals received by each array         element, or in inversion methods including pseudo-inversion         techniques used to jointly determine the beamformed signals. 

1. A method for transmitting digital beamformed signals in a transmit array antenna apparatus utilizing a single transceiver with one power amplifier, one up-frequency converter and one digital-to-analog converter for said array transmit antenna apparatus comprising the steps of: generating a first set of direct-sequence spread spectrum codes; generating a plurality of weights, each weight being a beamforming amplitude and phase or delay for each element; generating a direct-sequence spread spectrum multiplexed signal containing such weights while using one of such first-set codes per element; converting such an multiplexed signal to a convenient radio frequency; amplifying and transmitting such a multiplexed radio frequency signal to the elements; generating a second set of direct-sequence spread spectrum codes; extracting a radio frequency signal with direction-bearing weight information at each element while using a subset of codes from the second set; generating a third set of direct-sequence spread spectrum codes at each element; transmitting a signal with array gain beamformed towards a specific direction while using a transmit array apparatus composed of spaced elements, such a transmit beamformed signal being a radio frequency signal, a direct-sequence spread spectrum radio frequency signal containing a subset of codes from the third set, or a sequence of radio frequency pulses that have short duration and high power.
 2. A method for receiving direct-sequence spread spectrum signals containing a plurality of codes with digital beamforming in a receive array antenna apparatus utilizing a single transceiver with one low noise amplifier, one down-frequency converter and one analog-to-digital converter for said receive array antenna apparatus comprising the steps of: receiving a subset of codes of said direct-sequence spread spectrum signals from multiple positions or from one or multiple transmitters located at different positions while using a receive array apparatus composed of spaced-elements; reflecting such received signals at each element; combining such reflected signals forming a common aggregated signal for the array; converting such an aggregate signal to a convenient intermediary frequency including baseband; sampling such a frequency-converted signal; correlating resulting samples against codes or variations of such codes included in the reflected signals; demultiplexing the reflected signals; detecting direction-bearing parameters included in such demultiplexed signals including joint detection of such parameters and data included in the incident signals, generating a receive beamformed signal towards one or more selected directions included or not in such direction-bearing parameters, such a receive beamformed signal being a baseband signal with receive array gain towards such selected directions.
 3. The method of claim 1, wherein each code of the first-set codes is a row of a circulant matrix.
 4. The method of claim 1, wherein each row of the circulant matrix is a cyclic shift of a common maximal-length sequence.
 5. The method of claim 1, wherein each code of the first set is a scaled version of a row of a circulant matrix.
 6. The method of claim 1, wherein the at least one code of the first set is orthogonal to at least one code of the second set.
 7. The method of claim 1, wherein codes from different rows of first and second sets are mutually orthogonal.
 8. The method of claims 1, 4 and 7, wherein codes of first set have values ±1 and each code of the second set is obtained from a corresponding row from the first set by replacing the negative values (i.e., −1) by zeros.
 9. The method of claim 1, wherein the transmitting step includes the sub-steps of: modulating the information using a first modulation; combining the modulated information with one of the codes of the first set; converting the modulated information combined with the codes in a multi-level aggregate baseband signal, up-converting the baseband signal to a radio frequency signal, power-amplifying the radio frequency signal, and sending the amplified aggregate radio frequency signal to a subset including all elements of the array.
 10. The method of claim 9, wherein the first modulation is M-ary Phase-Shift Keying or PSK, including Binary Phase Shift Keying or BPSK.
 11. The method of claim 9, wherein the first modulation is M-ary Quadrature Amplitude Modulation or QAM including Quadrature-Phase Shift Keying or QPSK and Offset Quadrature-Phase Shift Keying or OQPSK.
 12. The method of claim 9, wherein the modulated symbol has an arbitrary complex value.
 13. The method of claim 12, wherein the complex values assigned to codes with consecutive cyclic shifts results from discrete time signal including samples from a continuous time signal including samples from a sensor device or system.
 14. The method of claim 12, wherein the complex values assigned to codes with consecutive cyclic shifts has a specified timing or spectral characteristic including radar signals with specified time and frequency resolution.
 15. The method of claims 1 and 9, wherein the sending of the aggregate radio frequency signal to the elements includes wireless and wired means.
 16. The method of claim 1, wherein the radio frequency signal for each element is extracted from the aggregate radio frequency signal transmitted to all the elements by correlating it against codes of the second set.
 17. The method of claims 1 and 16, wherein the correlation is implemented at each element by multiplying or re-modulating the aggregate signal with one or a sub-set of codes from the second set followed by filtering the resulting radio frequency signal by a narrow high-Q band-pass filter centered at the carrier frequency for the array.
 18. The method of claims 1 and 17, wherein the filtered radio frequency signal is sent to and transmitted by the antenna element without further re-modulation or conditioning.
 19. The method of claims 1 and 17, wherein the filtered radio frequency signal is re-modulated using one or a subset of codes from the third set, and then sent to and transmitted by the antenna element as a radio frequency spread spectrum signal.
 20. The method of claims 1, 18 and 19, wherein the resulting radio frequency signal (i.e., ‘as is’ or spread spectrum) is segmented in wave-blocks and each wave-block is compressed and transmitted as a radio frequency pulse.
 21. The method of claims 1, 19 and 20, wherein a wave-block corresponds to one or a fraction of a chip of the spread spectrum signal.
 22. The method of claims 1 and 20, wherein compression of a wave-block in a pulse is accomplished through coherent accumulation of segments of such a wave-block, each segment corresponding to a specified number of integer cycles at the carrier frequency for the array.
 23. The method of claims 1 and 22, wherein pulse power amplification results from incremental and recurring coherent energy accumulation as the radio frequency signal propagates over a delay line.
 24. The method of claims 1, 22 and 23, wherein the number of cycles per pulse is determined by the length of such delay line.
 25. The method of claims 1, 22 and 23, wherein the power amplification of the radio frequency pulse is determined by the relative lengths of the wave-block and delay line.
 26. The method of claims 1, 22 and 23, wherein the coherent power amplification is implement as in FIG. 16 using just passive components not including capacitors or inductances.
 27. The method of claims 1, 19 and 20, wherein the radio frequency signal transmitted beamformed in the direction of a receiver array is a direct-sequence spread spectrum radio frequency signal containing one or a subset of codes with reflective properties compatible with the receive beamforming methods included in this invention, including spread spectrum signals in which each chip is transmitted as one or a set of short-duration pulses.
 28. The method of claim 2 and 27, wherein each code included in the spread spectrum radio frequency signal incident in the receive array is a row of a circulant matrix.
 29. The method of claims 2 and 28, wherein each row of the circulant matrix is a different cyclic-shift of a common maximal-length sequence.
 30. The method of claims 2, 27 and 28, wherein each code of the incident radio frequency spread spectrum signal is a scaled version of a row of a circulant matrix.
 31. The method of claims 2, 27 and 28, wherein the signal incident on the array includes a plurality of spread spectrum signals.
 32. The method of claim 2, 27 and 28, wherein the signal incident on the array includes signals from a plurality of transmitters at difference locations.
 33. The method of claim 2 and 31, wherein at least one of the incident signals from at least one of the transmitters includes multipath reflections from urban structures such as mountains and buildings.
 34. The method of claim 2, 31 and 32, wherein at least one of the incident signals from at least one of the transmitter include reflections, including multipath reflections from a plurality of stationary or moving objects such as airplanes.
 35. The method of claim 2, wherein the reflecting signal of each element have distinguishable spectrum features including distinguishable amplitudes and delays.
 36. The method of claims 2, 28 and 35, wherein the reflecting signal is includes a row of the circulant matrix.
 37. The methods of claims 2, 28, 29 and 35, wherein the reflecting signal is circulating and does not include a maximal-length sequence included in the received signals.
 38. The method of claim 2, wherein the reflected signal by each element results from the multiplication or re-modulation of the incident signal at each element by the reflecting signal at such an element.
 39. The methods or claims 2 and 38, wherein the reflected signals are combined using a cable.
 40. The methods of claims 2 and 38, wherein the reflected signals are combined using a microwave signal combiner including a Wilkinson combiner.
 41. The methods of claims 2 and 38, wherein the reflected signals propagate away from the antenna elements through backscattering and combine wirelessly at a location remote from the receive elements.
 42. The method of claims 2 and 38, wherein the reflected signal at each element result from an impedance mismatch between the antenna element and the load circuit.
 43. The method of claim 42, wherein the reflecting signal at each element controls the impedance of such a load.
 44. The method of claims 2 and 38, wherein the reflected signals are generated using surface acoustic wave (SAW) devices.
 45. The method of claim 2, wherein a single digital receiver for the entire array is used to used to amplify down-convert and sample the aggregate signal resulting from the combination of reflected signals.
 46. The methods of claims 2, 38 and 45, wherein the reflecting signals are synchronized with the sample-and-hold devices included in the analog-to-digital converter such that sampling is performed half-way between consecutive transitions included in the reflecting signals as the resulting reflected signal are received at the analog-to-digital converter.
 47. The methods of claims 2 and 45, wherein the analog-to-digital converter operates at least at the Nyquist frequency of the aggregate signal resulting from the reflected signals while producing at least one complex-valued sample per chip time.
 48. The method of claim 2, wherein a correlator apparatus including software operations are used to demultiplex and detect beamforming-bearing information for each element and included in the aggregate signal that result from the combination of reflected signals.
 49. The methods of claims 2, 29 and 48, wherein the codes used in the correlation are derived from maximal length sequences included in the incident signals.
 50. The methods of claims 2 and 49, wherein the correlation is performed using select and add operations.
 51. The methods of claims 2, 48 and 49, wherein the correlation results from selecting just the sample positions corresponding to positive values of such cyclically shifted maximal length sequences.
 52. The method of claims 2 and 48, wherein direction-bearing information detected from at least one of the incident signals are used in the beamforming operations for isolating or nulling such at least one of the signals.
 53. The method of claim 2, wherein weights that include information of one, a set or all the incident angles are used in the beamforming operations for separating signals incident at different angles.
 54. The methods of claims 2, 52 and 53, wherein the beamforming operations include forming nulls toward one or a set of specified angular directions, including directions of at least one of the incident signals.
 55. The methods of claims 2, 52, 53 and 54, wherein the beamforming operations include joint detection of modulated symbols, complex-valued samples, complex-valued data and multipath-bearing information included in the incident signals. 